Apparatus and method for a dual output resonant converter to ensure full power range for both outputs

ABSTRACT

A power converter including: a dual output resonant converter including a first output, a second output, a common mode control input, and a differential mode control input, wherein a voltage/current at the first output and a voltage/current at the second output are controlled in response to a common mode control signal received at the common mode control input and a differential mode control signal received at the differential mode control input; and a dual output controller including a first error signal input, a second error signal input, a delta power signal input, a common mode control output, and a differential mode control output, wherein the dual output controller is configured to generate the common mode control signal and the differential mode control signal in response to a first error signal received at the first error signal input and a second error signal received at the second error signal input, wherein the first error signal is a function of the voltage/current at the first output and the second error signal is a function of the voltage/current at the second output, and wherein the common mode control signal is output from the common mode control output and the differential mode control signal is output from the differential mode control output, wherein the differential mode signal is limited by a differential mode signal limiting circuit.

TECHNICAL FIELD

Various exemplary embodiments disclosed herein relate generally to amethod and apparatus for linearization of the control inputs for a dualoutput resonant converter.

BACKGROUND

For power converters operating at powers larger than approximately 100watts (W) at full load, a resonant topology provides a solutions withhigh efficiency and small volumes/high power density. At power levelsabove 100 watts, the extra cost compared to other topologies (e.g., anextra switch, extra secondary diode, resonant capacitor) is compensatedfor by additional advantages of the resonant topology. There are severaltypes of resonant converters, such as a series resonant converter, anLLC converter, and an LCC converter. The series resonant converter usesa resonant capacitor, Cr, and an inductor, Ls, as resonating componentswhile LLC and LCC converters use three resonant components. For an LLCconverter, the magnetizing inductance of the transformer takes part inthe resonance, while for an LCC converter, an extra capacitor, whichtakes part in the resonance, is present at the secondary side of thetransformer.

Resonant power supplies are being used in light emitting diode (LED)television applications to provide a low voltage output of about 12V DCthat supplies the low voltage circuits and a high voltage output ofaround 165V that supplies the LED strings for the backlight of thedisplay. Such resonant power supplies typically include a mainregulation loop that senses the 12V output and regulates the power levelof the converter in order to keep the 12V output constant while the loadvaries. The 165V output then follows the 12V by a more or less fixedratio as set by the turns ratio of the resonant transformer. Because the165V output is not regulated, the output voltage of the 165V output canvary considerably with load variations on both the 165V output and the12V output. Thus, a second control stage is often used after the 165Voutput to provide for a more accurate supply voltage for the LEDstrings. However, a second control stage adds cost to such resonantpower supplies.

SUMMARY

A brief summary of various exemplary embodiments is presented below.Some simplifications and omissions may be made in the following summary,which is intended to highlight and introduce some aspects of the variousexemplary embodiments, but not to limit the scope of the invention.Detailed descriptions of an exemplary embodiment adequate to allow thoseof ordinary skill in the art to make and use the inventive concepts willfollow in later sections.

Various embodiments relate to a power converter including: a dual outputresonant converter including a first output, a second output, a commonmode control input, and a differential mode control input, wherein avoltage/current at the first output and a voltage/current at the secondoutput are controlled in response to a common mode control signalreceived at the common mode control input and a differential modecontrol signal received at the differential mode control input; and adual output controller including a first error signal input, a seconderror signal input, a delta power signal input, a common mode controloutput, and a differential mode control output, wherein the dual outputcontroller is configured to generate the common mode control signal andthe differential mode control signal in response to a first error signalreceived at the first error signal input and a second error signalreceived at the second error signal input, wherein the first errorsignal is a function of the voltage/current at the first output and thesecond error signal is a function of the voltage/current at the secondoutput, and wherein the common mode control signal is output from thecommon mode control output and the differential mode control signal isoutput from the differential mode control output, wherein thedifferential mode signal is limited by a differential mode signallimiting circuit.

Various embodiments are described, wherein the differential mode signallimiting circuit includes: a first maximum detector receiving a scaledfirst error signal at a first input and a first differential mode signallimit at a second input, wherein the first maximum detector isconfigured to produce an output that is the maximum value received atthe first input and the second input; a second maximum detectorreceiving a scaled second error signal at a third input and a seconddifferential mode signal limit at a fourth input, wherein the secondmaximum detector is configured to produce an output that is the maximumvalue received at the third input and the fourth input; a first adderconfigured to produce an output that is the addition of the firstmaximum detector output and the second maximum detector output; and asecond adder configured to produce the differential mode signal bysubtracting the output of the first adder from a differential modesignal maximum value Vdm_max.

Various embodiments are described, wherein the differential mode signalmaximum value is scaled by a value k wherein 1<k<1.1.

Various embodiments are described, wherein the first differential modesignal limit is calculated as Vdm_max*alpha/m, wherein alpha and m arescale factors such that 0<alpha<1 and 1<m<1.1.

Various embodiments are described, wherein the common mode controlsignal is generated using a feedback loop that uses a desired deltapower signal based upon the first error signal and the second errorsignal and a delta power signal that is a function of the difference inoutput power at the first output and the second output.

Various embodiments are described, wherein the dual output controller isconfigured to generate the common mode control signal and thedifferential mode control signal in response to the first error signaland the second error signal by precalculating a control variable matrixand generating the common mode control signal and the differential modecontrol signal as a function of the first and second error signals andthe control variable matrix.

Various embodiments are described, wherein the control variable matrixincludes variables k11, k12, k21, and k22, wherein the desired deltapower signal and the differential mode control signal are generated as:desired_delta_power=first error signal·k21+second error signal·k22; andVdm=first error signal·k11+second error signal·k12.

Various embodiments are described, wherein including a clamping circuitconfigured to clamp the common mode control signal to a range of values.

Various embodiments are described, wherein the clamping circuit furtherincludes: a power detector configured to produce an indication signalwhen the output power of the first output approaches zero; and a limitdetector configured to receive the common mode signal and the indicationsignal to produce a limited common mode signal based upon the range ofvalues.

Various embodiments are described, wherein the power detector includes areference input configured to receive a reference signal and avoltage/current input that receives a voltage/current signal from thefirst output, wherein the indication signal produced based upon thereference signal the voltage/current signal from the first output.

Various embodiments are described, wherein the power detector includesan auxiliary voltage input configured to receive an auxiliary voltagesignal from an auxiliary coil coupled to a secondary of the dual outputresonant converter, wherein the indication signal is produced based uponthe auxiliary voltage signal.

Various embodiments are described, wherein the power detector includesan auxiliary voltage input configured to receive an auxiliary voltagesignal from an auxiliary coil coupled to a secondary of the dual outputresonant converter and a switching node voltage input that receives aswitching mode voltage signal from a switching node in the dual outputresonant converter, wherein the indication signal is produced based uponthe auxiliary voltage signal and the switching node voltage signalindicating the beginning a continuous conduction mode (CCM) of operationof the dual output resonant converted.

Further various embodiments relate to a power converter including: adual output resonant converter including a first output, a secondoutput, a duty cycle control input, and a frequency control input,wherein a voltage/current at the first output and a voltage/current atthe second output are controlled in response to a duty cycle controlsignal received at the duty cycle control input and a frequency controlsignal received at the frequency control input; and a dual outputcontroller including a duty cycle control output and a frequency controloutput, wherein the dual output controller is configured to generate theduty cycle control signal and the frequency control signal in responseto the voltage/current at the first output and a voltage/current at thesecond output, wherein the duty cycle control signal is limited by aduty cycle control signal limiting circuit.

Various embodiments are described, wherein the duty cycle control signalcircuit includes: a first maximum detector receiving a scaled firsterror signal at a first input and a first duty cycle control signallimit at a second input, wherein the first maximum detector isconfigured to produce an output that is the maximum value received atthe first input and the second input; a second maximum detectorreceiving a scaled second error signal at a second input and a secondduty cycle control signal limit at a third input, wherein the secondmaximum detector is configured to produce an output that is the maximumvalue received at the third input and the fourth input; a first adderconfigured to produce an output that is the addition of the firstmaximum detector output and the second maximum detector output; and asecond adder configured to produce the duty cycle control signal bysubtracting the output of the first adder from a duty cycle controlsignal maximum value Vdc_max.

Various embodiments are described, wherein the duty cycle control signalmaximum value is scaled by a value k wherein 1<k<1.1.

Various embodiments are described, wherein the first duty cycle controlsignal limit is calculated as Vdc_max*alpha/m, wherein alpha and m arescale factors such that 0<alpha<1 and 1<m<1.1.

Various embodiments are described, wherein the duty cycle control signalis generated using a first feedback loop that uses a desired delta powersignal based upon the voltage/current at the first output and avoltage/current at the second output and a delta power signal that is afunction of the difference in output power at the first output and thesecond output and wherein the frequency signal is generated using asecond feedback loop that uses a desired total power signal based uponthe voltage/current at the first output and a voltage/current at thesecond output and a total power signal that is a function of the totalpower at the first output and the second output.

Various embodiments are described, further including a clamping circuitconfigured to clamp the duty cycle control signal to a range of values.

Various embodiments are described, further including a clamping circuitconfigured to clamp the frequency control signal to a range of values.

BRIEF DESCRIPTION OF THE DRAWINGS

In order to better understand various exemplary embodiments, referenceis made to the accompanying drawings, wherein:

FIG. 1 depicts and embodiment of a dual output resonant converter;

FIG. 2 is a block diagram of an example of a power converter thatincludes a dual output resonant converter, a dual output controller, andfirst and second compare units in accordance with an embodiment of theinvention;

FIG. 3 is a graph of simulation results for a power converter thatincludes a dual output resonant converter;

FIG. 4 is a table of some of the data points in the graph of FIG. 3along with some corresponding partial derivative values;

FIG. 5 illustrates an example of the control function that is performedby the dual output controller.

FIG. 6A is a block diagram of an example of a power converter that isconfigured for Vcap or Vcr control;

FIG. 6B depicts an embodiment of a dual output resonant converter thatis configured for Vcap control;

FIG. 6C is a block diagram of an example of a power converter that isconfigured for Vcr control;

FIGS. 7A and 7B illustrate plots of Vcm and Vdm versus the powersupplied to an LED (Pled) string for various LED voltage levels;

FIG. 8 illustrates embodiment of a dual output controller for a dualoutput resonant converter that linearizes Vcm;

FIG. 9 illustrates the relationship between

Ec_desired and Vcm as result of the feedback loop;

FIG. 10 illustrates the linear relationship between

Ec_desired and Pled;

FIG. 11 illustrates dual output resonant converter 1100 the usingfeedback loops to linearize the duty cycle and frequency controlparameters;

FIG. 12 illustrates a block diagram of a first embodiment for clampingVcm;

FIG. 13 illustrates a block diagram of a second embodiment for clampingVcm;

FIG. 14 illustrates a block diagram of the aux voltage detector;

FIG. 15 illustrates a plot of current on the primary coil and in thecore in the situation when both power outputs are loaded;

FIG. 16 illustrates a plot of current on the primary coil and in thecore in the situation when one output has gone to zero output power;

FIG. 17 illustrates a block diagram of a third embodiment for clampingVcm;

FIG. 18 illustrates a block diagram of the CCM detector;

FIG. 19 illustrates a simulation result of a dual output resonantconverter being controlled by Vdm and Vcm signals;

FIG. 20 illustrates a simplified model of the plot of FIG. 19;

FIG. 21 illustrates a block diagram of a dual output controller showingthat when the measured currents are zero so that a value of Vdm_max isoutput from the controller; and

FIG. 22A illustrates the situation where Vcm is no longer capable ofsetting the division of power between both outputs;

FIG. 22B illustrates an example of the dual output resonant converterbeing controlled by Vdm and Vcm;

FIG. 23 illustrates an embodiment of a dual output controller thatlimits the value of Vdm;

FIG. 24 illustrates a first embodiment of a dual output controller usinga Vcm offset;

FIG. 25 illustrates a second embodiment of a dual output controllerusing a Vcm offset;

FIG. 26 illustrates an embodiment of a Vcm_offset circuit;

FIG. 27 illustrates an embodiment of a converter that limits the errorsignals;

FIG. 28 illustrates an embodiment of zero output power processor;

FIG. 29 illustrates an embodiment of zero output power processor basedupon slowly adapting the level of Vcmoffset; and

FIG. 30 illustrates an embodiment of a converter that limits the errorsignals using duty cycle and frequency controls.

To facilitate understanding, identical reference numerals have been usedto designate elements having substantially the same or similar structureand/or substantially the same or similar function.

DETAILED DESCRIPTION

The description and drawings illustrate the principles of the invention.It will thus be appreciated that those skilled in the art will be ableto devise various arrangements that, although not explicitly describedor shown herein, embody the principles of the invention and are includedwithin its scope. Furthermore, all examples recited herein areprincipally intended expressly to be for pedagogical purposes to aid thereader in understanding the principles of the invention and the conceptscontributed by the inventor(s) to furthering the art, and are to beconstrued as being without limitation to such specifically recitedexamples and conditions. Additionally, the term, “or,” as used herein,refers to a non-exclusive or (i.e., and/or), unless otherwise indicated(e.g., “or else” or “or in the alternative”). Also, the variousembodiments described herein are not necessarily mutually exclusive, assome embodiments can be combined with one or more other embodiments toform new embodiments.

Dual output resonant converters are known. Examples of dual outputresonant converters are disclosed in U.S. Pat. Nos. 6,822,881 and6,829,151. An embodiment of a dual output resonant converter 100 isdepicted in FIG. 1. In the embodiment of FIG. 1, the circuit is aresonant LLC converter that includes a resonant capacitor, Cr, andinductor, Ls, and magnetizing inductance, Lm, which are components thatform part of a resonant circuit or a resonant tank. Transformer andrectifier circuits are used to create DC output voltages, Vout1 andVout2. The output current can be made continuous by adding a seriesinductance (not shown). In an embodiment of FIG. 1, the circuit includesthree parts. A first part 1 is the control part, which includes controllogic, 5, for generating control signals for opening and closing theswitches, 6 and 7, by means of a high-side driver (HSD) 8 and a low-sidedriver (LSD) 9, respectively. A second part 2 is the primary circuit anda third part 3 is the secondary circuit. The second part includes theresonant capacitor, Cr, and the inductors, Ls and Lm. The resonantconverter is connected to a voltage supply, Vsupply, (also referred toherein as Vbus) so that electrical energy may be supplied to a loadthat, for example, may be connected to output terminals at the secondaryside. In an embodiment, the switches, 6 and 7, are series-arranged,controllable switches that are connected to the voltage supply, Vsupply,the first switch being a high-side switch (HSS), the high-side switchbeing connected at one leg to the voltage supply, Vsupply, the secondswitch being a low-side switch (LSS), the low-side switch beingconnected at one leg to ground. Also, an embodiment with a full bridgeconfiguration can be envisioned. The third part may include diodes andcapacitors as is known in the field.

The dual output resonant converter 100 is typically controlled by afrequency. The output power of the converter may be controlled byvarying the signal frequency. It is also possible to use a duty cycledifferent from 50%. In this case, the duty cycle also influences how theoutput power is divided between both outputs. A drawback to frequencyand duty cycle based control may be that for certain operating points itbecomes difficult to regulate the system to the desired operating pointbecause of changes in gain factors due to nonlinear control behavior ofa frequency or duty cycle controlled resonant converter and even changesin polarity.

Various techniques for operating resonant converters are known. Onetechnique as disclosed in U.S. Pat. No. 7,944,716, which is incorporatedby reference herein for all purposes as if fully set forth herein,involves combining control of the duty cycle and the frequency in such away that a smooth regulation of the output power can be realized. Forexample, the resonant converter is not controlled by frequency and dutycycle directly, but with current and voltage at the primary side of thetransformer. In an embodiment, current and voltage in the resonant tank,e.g., Iprim and Vcap, are compared each conduction interval with twocontrol values such that the resonant converter is controlled in acycle-by-cycle manner. In an embodiment, the current, Iprim, is thecurrent flowing in the resonant tank in response to the opening andclosing of the switches. Measurement of the current may be provided invarious different ways, e.g., from the voltage over a sense resistor,the current in a switch, etc. The current, Iprim, is also referred to asthe primary current. Vcap is also referred to as the capacitor voltage,which is the voltage at a predetermined point, such as the nodeindicated as “Vcap” in FIG. 1. In the example of FIG. 1, the voltage atnode Vcap is defined as Vhb−Vcr, where Vhb is the voltage at the halfbridge node 11 and Vcr is the voltage across the resonant capacitor. Inan example operation, for each half cycle, the conducting primary switchis switched off when the voltage, Vcap, crosses a predefined value foreach half cycle (e.g., VcapH for high-side switch turn off and VcapL forlow-side switch turn off). By controlling the primary switches in thisway, it is possible to get an almost linear relationship between acontrol parameter and output power.

Further, it is possible to define a differential mode term, Vdm, and acommon mode term, Vcm, for use in controlling a resonant converter. Forexample, the differential mode term may be expressed asVdm=Vbus/2−(VcapH−VcapL)/2 and the common mode term may be expressed asVcm=(VcapH+VcapL)/2. Using the differential mode term and the commonmode term, it is possible to control both the total power that isdelivered to the output using the differential mode term, Vdm, while thedifference in output power that is delivered to the output during eachhalf cycle is determined by the common mode term, Vcm.

It is possible to use the voltage across a resonant capacitor, Vcr, orthe voltage at the node Vcap to implement a control scheme that utilizesa differential mode term and a common mode term. The techniquesinvolving Vcap and Vcr can yield similar control functionality. However,a difference between using Vcap control versus Vcr control is that theaverage value of Vcap is per definition zero, while for Vcr, a DCcomponent, Vbus/2, is present, which makes it possible to use acapacitive divider to divide Vcap down to a low voltage signal that iscompatible with a low voltage controller integrated circuit (IC). Whilethe DC component is Vbus/2 for Vcr control at a 50% duty cycle, for dutycycles other than 50%, the DC component is Vbus/2×duty cycle, where theduty cycle is the high-side switch conduction time divided by the periodtime.

Additional techniques for operating resonant converters that use adifferential mode term, Vdm, and a common mode term, Vcm, to control aresonant converter are disclosed in U.S. Pat. No. 9,065,350, andEuropean Patent Application No. 11250662.1 (Published as EP 2 547 176A1, on Jan. 16, 2013), which are incorporated by reference herein forall purposes as if fully set forth herein.

In addition to the 50% duty cycle modes or “high power modes,” it isalso known that it is possible to use “low power modes” to control aresonant converter. Various examples of low power modes for controllingresonant converters are disclosed in for example, U.S. Pat. No.8,339,817, International Patent Applications Published under the PatentCooperation Treaty (PCT) as publication numbers WO 2005/112238 and WO2009/004582, and European Patent Application No. 15159086.6 (Publishedas EP 3 068 027 A1, on Mar. 13, 2015). In such low power modes, part ofa switching sequence is similar to the high power mode, while duringanother part of the cycle, the resonant tank is in a mode where noenergy is converted and where also relatively few losses are produced.

As described above, resonant power supplies are being used in LEDtelevision applications to provide a low voltage output of about 12V DCthat supplies the low voltage circuits and a high voltage output ofaround 165V that supplies the LED strings for the backlight of thedisplay. Some known power supplies developed for such applicationsrequire a second control stage after the 165V output to provide for amore accurate supply voltage for the LED strings. However, a secondcontrol stage adds cost to such resonant power supplies. In accordancewith an embodiment described herein, a power converter with a dualoutput resonant converter is disclosed that does not need a second stagecontroller for the 165V output. The power converter is controlled usinga “capacitor voltage” control technique so that the two outputs of thedual output resonant converter can be controlled independently of eachother. In an embodiment, a differential mode control signal, Vdm, and acommon mode control signal, Vcm, are used to independently control thetwo outputs of a dual output resonant converter. In particular, thedifferential mode control signal, Vdm, and the common mode controlsignal, Vcm, are adjusted in response to error signals that aregenerated as a function of the output voltage and/or current of each ofthe two outputs. For example, a first (e.g., linear) combination of thefirst and second error signals is used to generate a differential modecontrol signal and a second (e.g., linear) combination of the first andsecond error signals is used to generate a common mode control signal.It is noted that the use of the dual output resonant converter in an LEDTV application is described, the embodiments described herein may beused in any application using a dual output resonant converter.

A technique for controlling a dual output resonant converter is firstdescribed with reference to FIG. 2. FIG. 2 is a block diagram of anexample of a power converter 200 that includes a dual output resonantconverter 210, a dual output controller 220, and first and secondcompare units 230 and 232 in accordance with an embodiment. The dualoutput resonant converter 210, the first and second compare units 230and 232, and the dual output controller 220 form a feedback loop that isused to independently control the voltage and/or current at the twooutputs of the dual output resonant converter. For example, in an LEDtelevision application, the outputs can be independently controlled toprovide a 12V output and a 165V output (e.g., within ±10% or within ±5%of the target output) that can be used to drive the different componentsof an LED television. The dual output resonant converter includes acommon mode control input 212, a differential mode control input 214, afirst output 216, and a second output 218. The common mode control input212 receives a common mode control signal, Vcm, and the differentialmode control input 214 receives a differential mode control signal, Vdm,while the first output 216 provides an output voltage, Vout1, and thesecond output 218 provides an output voltage, Vout2. It should be notedthat the outputs could also be viewed in terms of an output current,Iout1 and Iout2, respectively.

The compare units 230 and 232 are configured to compare an input voltageto a reference voltage to generate an error signal that can be processedby the dual output controller 220. In an embodiment, the error signalsreflect the difference between the output voltage and a referencevoltage. For example, the compare units are used so that the 12V and165V outputs can be transformed to lower voltage signals that can bemanaged by an IC-based dual output controller, which typically operatesat voltages in the range of about 0.5-3 volts. The first compare unit230 is configured to compare the output, Vout1, from the first output216 to a first reference signal, Vref1, to generate a first errorsignal, error1, and the second compare unit 232 is configured to comparethe output, Vout2, from the second output 218 to a second referencesignal, Vref2, to generate a second error signal, error2. Although thecompare units and corresponding signals are described in terms ofvoltage, the compare units 230 and 232 may be configured as currentcompare units with corresponding current based signals, e.g., Iout1,Iout2, Iref1, and Iref2.

The dual output controller 220 includes a first error signal input 222,a second error signal input 224, a common mode control output 226, and adifferential mode control output 228. The first error signal input 222receives the first error signal, error1, and the second error signalinput 224 receives the second error signal, error2. The common modecontrol output 226 outputs the common mode control signal, Vcm, and thedifferential mode control output 228 outputs the differential modecontrol signal, Vdm.

A power converter with a dual output resonant converter may becontrolled based on the voltage at the node, Vcap, which is referred toherein as “Vcap control” or based on the voltage across the resonantcapacitor, which is referred to herein as “Vcr control.” In anembodiment, the voltage across the resonant capacitor, Vcr, may alsoinclude a voltage drop across a sense resistor, Rsense, but as the senseresistor is typically only in the 100 mV range while Vcr is in a rangethat is greater than 100V, the voltage drop attributable to the sensorresistor is insignificant. Depending on the placement of the resonantcapacitor, e.g., in series with the switching node or in series with theground node, a different signal shape occurs, so the common mode controlsignal, Vcm, and the differential mode control signal, Vdm, are definedslightly different between both configurations, but the basic principleapplied for power converter control is the same. For example, for Vcapcontrol:VcapH=Vbus/2+Vcm−Vdm;VcapL=−Vbus/2+Vcm+Vdm;

or written in another way;

-   -   Vdm=Vbus/2−(VcapH−VcapL)/2 controls Pout1+Pout2 with only a        small residual effect on Pout1−Pout2;    -   Vcm=(VcapH+VcapL)/2 controls Pout1−Pout2 with only a small        residual effect on Pout1+Pout2;    -   where Pout1 is the power delivered on the first output 216 of        the dual output resonant converter 210 and Pout2 is the power        delivered on the second output 218 of the dual output resonant        converter.

For Vcr control compared to Vcap control, a one-to-one relationshipexists between values, e.g., for every value of VcapH and VcapL onevalue exists for VcrH and VcrL. For example, for Vcr control:Vhb=Vcap+Vcr so;

-   -   VcapH level is relevant when the high-side switch is on, then        Vhb=Vbus so Vcr=Vhb−Vcap gives:        VcrH=Vbus−VcapH;    -   VcapL level is relevant when low-side switch is on, then Vhb=0        so Vcr=Vhb−Vcap gives:        VcrL=−VcapL;        Vcm=(VcapH+VcapL)/2=((Vbus−VcrH)+(−VcrL))/2=Vbus/2−(VcrH+VcrL)/2.

Using Vcap control, the common mode control signal, Vcm, does not dependon Vbus, while with Vcr control, the common mode control signal doesdepend on Vbus, e.g., Vbus/2. Thus, in an embodiment that uses Vcapcontrol, the differential mode control signal is defined asVdm=Vbus/2−(VcapH−VcapL)/2 and the common mode term is defined asVcm=(VcapH+VcapL)/2 and in an embodiment that uses Vcr control, thedifferential mode control signal is defined as Vdm=(VcrH−VcrL)/2 and thecommon mode term is defined as Vcm=Vbus/2−(VcrH+VcrL)/2. As is describedbelow, in an embodiment, the dual output controller functions the samefor both Vcap control and Vcr control.

A consideration in choosing to use Vcap control or Vcr control isrelated to the specific implementation. For example, with Vcap control,the average voltage of Vcap is zero, Vcap=0 (because the voltage issensed across an inductor). Therefore, it is easy to use a capacitivedivider to translate the amplitude of Vcap (e.g., a few 100V) to IClevels of, for example, a few volts. A capacitive divider cannottransfer DC information, but because the DC information is zero perdefinition, it is also not required. In an embodiment, the DC term canbe set to zero, for example, by connecting a large resistor between thecapacitive divider and the ground reference that VcapH and VcapL arereferenced to. Vcr control may be selected so that a resonant capacitorcan be connected to ground at one side, which enables the resonantcapacitor to be split into two capacitors, which may give less ripplecurrent in the supply voltage, Vsupply, also referred to as Vbus.

As described above, the differential mode control signal, Vdm, and acommon mode control signal, Vcm, are used to independently control thetwo outputs of a dual output resonant converter. In particular, thedifferential mode control signal and the common mode control signal areadjusted in response to error signals, error1 and error2, that aregenerated as a function of the output voltage and/or current,Vout1/Iout1 and Vout2/Iout2, of each of the two outputs. An example of atechnique for deriving the functional relationships between Vcm, Vdm,error1, and error2 is described below.

In the power converter 200 depicted in FIG. 2, the error signalgenerated for one of the outputs should drive the combination of Vcm andVdm in the proper ratio such that only power to the desired outputchanges and the power to the other output stays constant. For example,the error signal corresponding to the first output, error1, should drivechanges to the output at the first output, Vout1, of the dual outputresonant converter with little effect (e.g., ±1% change) on the outputat the second output, Vout2, of the dual output resonant converter whilethe error signal corresponding to the second output, error2, shoulddrive the changes to the output at the second output, Vout2, of the dualoutput resonant converter with little effect on the first output, Vout1,of the dual output resonant converter. Such a control scheme is referredto herein as “orthogonal” control. In order to achieve orthogonalcontrol, in an embodiment, a first step is to describe how the outputcurrent (e.g., Iout1 and Iout2) at each output changes with changes inVcm and Vdm. The change in output current at each output as a functionof changes in Vcm and Vdm can be described as the total differential ofeach output based on partial derivatives as follows:dIout1=δiout1_dVcm·dVcm+δiout1_dVdm·dVdmdIout2=δiout2_dVcm·dVcm+δiout2_dVdm·dVdm

These partial derivatives can then be determined for a particularconfiguration of a dual output resonant converter operating over a setof known operating points. FIG. 3 is a graph of simulation results for apower converter that includes a dual output resonant converter asdescribed above with reference to FIGS. 1 and 2. In the example of FIG.3, the output current at two outputs, Iout1 and −Iout2, is plotted fordifferent values of the common mode control signal, Vcm, and fordifferent values of the differential mode control signal, Vdm. In theexample of FIG. 3, Vcm and Vdm correspond to the signal derived from theVcap or Vcr node by a voltage divider in order to get voltages at ascale applicable to an IC input. For example, the output currents, Iout1and −Iout2, are plotted over a range of common mode control signals,Vcm, from 0-0.6V and over a range of differential mode controls signals,Vdm, from 0-1.4V. FIG. 4 is a table of some of the data points in thegraph of FIG. 3 along with some corresponding partial derivative values.

In an embodiment, it is desired to determine the change in thedifferential mode control signal, Vdm, and the change in the common modecontrol signal, Vcm, needed in order to get a certain output currentchange at only one output (e.g., either Iout1 or Iout2), while leavingthe output current at the other output unchanged, e.g., unchanged withina range of about ±1% of full power of the output. In an embodiment, thedifferential mode control signal, Vdm, and the common mode controlsignal, Vcm, can be solved from the following set of equations:dIout1=δiout1_dVcm·dVcm+δiout1_dVdm·dVdmdIout2=δiout2_dVcm·dVcm+δiout2_dVdm·dVdmWhich gives, the matrix operation:

$\begin{pmatrix}{d\;{Iout}\; 1} \\{d\;{Iout}\; 2}\end{pmatrix} = {\begin{pmatrix}{{\delta i}\;{out}\; 1{\_ dVcm}} & {{\delta i}\;{out}\; 1{\_ dVdm}} \\{{\delta i}\;{out}\; 2{\_ dVcm}} & {{\delta i}\;{out}\; 2{\_ dVdm}}\end{pmatrix} \cdot \begin{pmatrix}{d\;{Vcm}} \\{d\;{Vdm}}\end{pmatrix}}$

Using the data in the example of FIGS. 3 and 4, at the operating pointVcm=0 and Vdm=0.4, the derivatives are determined as:δiout1_dVcm=45.28 δiout1_dVdm=25.7δiout2_dVcm=−47.5 δiout2_dVdm=24.14

The above values can then be applied to the equations below:

${d\;{Vcm}} = \frac{\begin{pmatrix}{d\;{Iout}\; 1} & {\delta\;{out}\; 1{\_ dVdm}} \\{d\;{iout}\; 2} & {\delta\;{out}\; 2{\_ dVdm}}\end{pmatrix}}{\begin{pmatrix}{\delta\;{iout}\; 1{\_ dVcm}} & {\delta\;{i{out}}\; 1{\_ dVdm}} \\{\delta\;{iout}\; 2{\_ dVcm}} & {\delta\;{i{out}}\; 2{\_ dVdm}}\end{pmatrix}}$ and ${d\;{Vdm}} = \frac{\begin{pmatrix}{{\delta out}\; 1{\_ dVcm}} & {d\; I\;{out}\; 1} \\{{\delta iout}\; 2{\_ dVcm}} & {d\; I\;{out}\; 2}\end{pmatrix}}{\begin{pmatrix}{\delta\;{iout}\; 1{\_ dVcm}} & {\delta\;{i{out}}\; 1{\_ dVdm}} \\{\delta\;{iout}\; 2{\_ dVcm}} & {\delta\;{i{out}}\; 2{\_ dVdm}}\end{pmatrix}}$

In an example, the changes needed in the values of the differential modecontrol signal, Vdm, and the common mode control signal, Vcm, to achievean output current change of 1 amp in each output can be determined fromthe above equations being a function of (dIout1,dIout2). For example,the changes needed in the values of Vdm and Vcm are calculated as:dVcm(1,0)=−11.107 ml dVcm(0,1)=10.433 ml dVcm(1,1)=−0.674 mldVdm(1,0)=19.569 ml dVdm(0,1)=20.529 ml dVdm(1,1)=40.098 ml

An example control function of an embodiment of the dual outputcontroller 220 is illustrated in FIG. 5. As illustrated in FIG. 5, thecommon mode control signal, Vcm, and the differential mode controlsignal, Vdm, are generated as a function of the input error signals,error1 (monitored as dIout1) and error2 (monitored as dIout2). Thevalues of the control parameters can be solved for and referred to ingeneral as control parameters k11, k12, k21, and k22. The controlparameters can be represented in a control parameter matrix of:

$\quad\begin{pmatrix}{k\; 11} & {k\; 12} \\{k\; 21} & {k\; 22}\end{pmatrix}$

With reference to FIG. 5, a determinant is used to solve the set oflinear equations in a structural way, e.g., as the quotient of twodeterminants. The determinant can be expressed as:

$\det = {{\begin{pmatrix}{\delta\;{iout}\; 1{\_ dVcm}} & {\delta\;{i{out}}\; 1{\_ dVdm}} \\{\delta\;{iout}\; 2{\_ dVcm}} & {\delta\;{i{out}}\; 2{\_ dVdm}}\end{pmatrix}} = {{\delta\;{iout}\; 1{{\_ dVcm} \cdot \delta}\;{iout}\; 2{\_ dVdm}} - {{\delta iout}\; 1{{\_ dVdm} \cdot {\delta{iout}2dVcm}}}}}$

And the set of equations can be solved as:

${{dVcm}\left( {{{dIout}\; 2},{{dIout}\; 1}} \right)} = \frac{{{dIout}\;{2 \cdot {\delta iout}}\; 1{\_ dVcm}} - \mspace{11mu}{{dIout}\;{1 \cdot {\delta iout}}\; 2{\_ dVcm}}}{{{\delta iout}\; 1{{\_ dVcm} \cdot {\delta iout}}\; 2{\_ dVdm}} - \mspace{11mu}{{\delta iout}\; 1{{\_ dVdm} \cdot {\delta iout}}\; 2{\_ dVcm}}}$${{dVdm}\left( {{{dIout}\; 2},{{dIout}\; 1}} \right)} = \frac{{{dIout}\;{1 \cdot {\delta iout}}\; 2{\_ dVdm}} - \;{{dIout}\;{2 \cdot {\delta iout}}\; 1{\_ dVdm}}}{{{\delta iout}\; 1{{\_ dVcm} \cdot {\delta iout}}\; 2{\_ dVdm}} - \;{{\delta iout}\; 1{{\_ dVdm} \cdot {\delta iout}}\; 2{\_ dVcm}}}$

Given the simulated values identified above:δiout1_dVcm=45.28δiout1_dVdm=25.7δiout2_dVcm=−47.5δiout2_dVdm=24.14

The control parameters k11, k12, k21, and k22 for the particularconfiguration of the dual output resonant converter are precalculatedas:

${k\; 11} = \frac{\delta\; i\;{out}\; 2{\_ dVdm}}{\det}$ k 11 = 0.011${k\; 12} = \frac{{- \delta}\; i\;{out}\; 1{\_ dVdm}}{\det}$k 12 = −0.011${k\; 21} = \frac{{- \delta}\; i\;{out}\; 2{\_ dVcm}}{\det}$k 21 = 0.019 ${k\; 22} = \frac{\delta\; i\;{out}\; 1{\_ dVcm}}{\det}$k 22 = 0.019

As shown above, a set of control parameters may be precalculated for aparticular configuration of a dual output resonant converter that isoperated and/or simulated over a known set of operating points. In anembodiment, the control parameters k11 and k12 are used by the dualoutput controller to set the common mode control signal, Vcm, and thecontrol parameters k21 and k22 are used by the dual output controller toset the differential mode control signal, Vdm. In an embodiment, forsymmetry reasons, when adapting the feedback loop starting from asymmetrical operating point where both outputs are equally loaded, k11and k12 should be opposite, while k21 and k22 should be equal. Althoughcertain values for the control parameters k11, k12, k21, and k22 arefound for an example power converter and an example set of operatingpoints, it should be understood that the particular values of thecontrol parameters are implementation specific. With the values of thecontrol parameters predetermined, the values of the common mode controlsignal, Vcm, and the differential mode control signal, Vdm, can begenerated using relatively simple calculations based on the errorvalues, error1 and error2. Thus, the two outputs of the dual outputresonant converter can be independently controlled in a feedback controlloop that utilizes two inputs and very little additional controlcircuitry.

FIG. 6A is a block diagram of an example of a power converter 300 thatis configured for Vcap control. The power converter can be similar to orthe same as the power converter 200 of FIG. 2 and includes a dual outputresonant converter 310, a dual output controller 320, and first andsecond compare units 330 and 332 in accordance with an embodiment of theinvention. The dual output resonant converter 310, the first and secondcompare units 330 and 332, and the dual output controller 320 form afeedback loop as described above with reference to FIG. 2. The dualoutput resonant converter 310 includes a common mode control input 312,a differential mode control input 314, a first output 316, and a secondoutput 318. The common mode control input 312 receives a common modecontrol signal, Vcm, and the differential mode control input 314receives a differential mode control signal, Vdm, while the first output316 provides an output voltage, Vout1, and the second output 318provides an output voltage, Vout2. It should be noted that the outputscould also be viewed in terms of an output current, Iout1 and Iout2,respectively.

The compare units 330 and 332 are configured to compare an input voltageto a reference voltage to generate an error signal. The first compareunit 330 is configured to compare the output, Vout1, from the firstoutput to a first reference signal, Vref1, to generate a first errorsignal, error1, and the second compare unit 332 is configured to comparethe output, Vout2, from the second output to a second reference signal,Vref2, to generate a second error signal, error2. Although the compareunits 330 and 332 and corresponding signals are described in terms ofvoltage, the compare units 330 and 332 could be configured as currentcompare units, with corresponding current based signals or as powerbased compare units for power-based regulation of the power converter.

The dual output controller 320 includes a first error signal input 322,a second error signal input 324, a common mode control output 326, and adifferential mode control output 328. The first error signal input 322receives the first error signal, error1, and the second error signalinput 324 receives the second error signal, error2. The common modecontrol output 326 outputs the common mode control signal, Vcm, and thedifferential mode control output 328 outputs the differential modecontrol signal, Vdm. As illustrated in FIG. 6A, the dual outputcontroller 320 generates the common mode control signal, Vcm, and thedifferential mode control signal, Vdm, in response to the error signals,error1 and error2. To generate the common mode control signal, Vcm, andthe differential mode control signal, Vdm, the dual output controller320 is configured with a control parameter matrix of:

$\quad\begin{pmatrix}{k\; 11} & {k\; 12} \\{k\; 21} & {k\; 22}\end{pmatrix}$and the control function can be expressed as:

$\begin{pmatrix}{Vdm} \\{Vcm}\end{pmatrix} = {\begin{pmatrix}{k\; 11} & {k\; 12} \\{k\; 21} & {k\; 22}\end{pmatrix} \cdot \begin{pmatrix}{{error}\; 1} \\{{error}\; 2}\end{pmatrix}}$

The control parameter matrix, which was described above, includes theparameters k11, k12, k21, and k22. The common mode control signal, Vcm,and the differential mode control signal, Vdm, can be generated as:Vcm=error1·k21+error2·k22; andVdm=error1·k11+error2·k12.

Thus, the control parameters k11-k22 define how the differential modecontrol signal, Vdm, and the common mode control signal, Vcm, change inresponse to changes in the error signals, error1 and error2. In anembodiment, the generation of the error signals, error1 and error2,includes an amplifier with frequency dependent behavior. Ultimately, thefunction is the specific linear combination of Vcm and Vdm to achieveorthogonal control. As described above, the relationship between controlparameters k11 and k21 determines how Vcm and Vdm change in response tochanges in the error signal, error1, such that a response is seen onlyat the first output, e.g., as Vout1/Iout1, and the relationship betweencontrol parameters k12 and k22 determines how Vcm and Vdm change inresponse to changes in the error signal, error2, such that a response isonly seen at the second output, e.g., as Vout2/Iout2.

FIG. 6B depicts an embodiment of a dual output resonant converter 410that is configured for Vcap control. The dual output resonant converter410 is similar to the dual output resonant converter described withreference to FIG. 1. However, the dual output resonant converter shownin FIG. 6B includes a controller 440 that is configured to generateswitch control signals in response to the common mode control signal,Vcm, the differential mode control signal, Vdm, and the voltage at node442. As shown in FIG. 6B, node 442 is identified as the “Vcap” node,e.g., the node at which Vcap is measured. In the circuit of FIG. 6B, thevoltage at Vcap can be expressed as: Vcap=Vhb−Vcr, where Vhb is thevoltage at the half bridge node 11 and Vcr is the voltage across theresonant capacitor, Cr. In operation, the output, Vout1, at the firstoutput 416 and the output, Vout2, at the second output 418 are fed backto the dual output controller through the compare units as describedabove with reference to FIGS. 2 and 6A. The common mode control signal,Vcm, and the differential mode control signal, Vdm, are generated by thedual output controller in response to the error signals, error1 anderror2. The common mode control signal, Vcm, and the differential modecontrol signal, Vdm, are provided to the controller of the dual outputresonant converter 410 and used to generate switch control signals thatare used by the control logic 5 to control the switching of thehigh-side and low-side switches 6 and 7 such that the output voltage,Vout1 and Vout2, at the first and second outputs, respectively, arecontrolled independent of each other.

The power converter 300 shown in FIG. 6A can also be configured for Vcrcontrol. When operating according to Vcr control, the dual outputcontroller 320 generates the common mode control signal, Vcr_cm, and thedifferential mode control signal, Vcr_dm, in response to the errorsignals, error1 and error2. To generate the common mode control signal,Vcr_cm, and the differential mode control signal, Vcr_dm, the dualoutput controller is configured with a control parameter matrix of:

$\quad\begin{pmatrix}{k\; 11} & {k\; 12} \\{k\; 21} & {k\; 22}\end{pmatrix}$and the control function can be expressed as:

$\begin{pmatrix}{Vcr\_ dm} \\{Vcr\_ cm}\end{pmatrix} = {\begin{pmatrix}{k\; 11} & {k\; 12} \\{k\; 21} & {k\; 22}\end{pmatrix} \cdot \begin{pmatrix}{{error}\; 1} \\{{error}\; 2}\end{pmatrix}}$

The control parameter matrix, which was described above, includes theparameters k11, k12, k21, and k22. The common mode term, Vcr_cm, and thedifferential mode term, Vcr_dm, are generated as:Vcr_cm=error1·k21+error2·k22; andVcr_dm=error1·k11+error2·k12.

FIG. 6C depicts an embodiment of a dual output resonant converter 510that is configured for Vcr control. The dual output resonant converter510 is similar to the dual output resonant converter described withreference to FIG. 1. However, the dual output resonant converter shownin FIG. 6C includes a controller 540 that is configured to generateswitch control signals in response to the common mode control signal,Vcr_cm, the differential mode control signal, Vcr_dm, and the voltageacross the resonant capacitor, Vcr. As shown in FIG. 6C, the voltage ismeasured across the resonant capacitor, Cr, and is identified as Vcr. Inoperation, the output, Vout1, at the first output 516 and the output,Vout2, at the second output 518 are fed back to the dual outputcontroller through the compare units as described above with referenceto FIGS. 2 and 6A. The common mode control signal, Vcr_cm, and thedifferential mode control signal, Vcr_dm, are generated by the dualoutput controller in response to the error signals, error1 and error2.The common mode control signal, Vcr_cm, and the differential modecontrol signal, Vcr_dm, are provided to the controller of the dualoutput resonant converter and used to generate switch control signalsthat are used by the control logic 5 to control the switching of thehigh-side and low-side switches 6 and 7 such that the output voltage,Vout1 and Vout2, at the first and second outputs, respectively, arecontrolled independent of each other.

The dual output controller basically does the inverse action of the dualoutput resonant converter such that a change in the error1 signal onlygives a change in the power delivered to Vout1, while a change in theerror 2 signal only gives a change in the power delivered to Vout2. Asdiscussed above, this method of control is called orthogonal controlwhich means that the power in both outputs can be changed independentlyof each other. Compared to the frequency controlled resonant converterof U.S. Pat. Nos. 6,822,881 and 6,829,151B2, the control by statevariables gives an almost linear relation between Vdm control input andoutput power. This makes it easier to get the desired orthogonalcontrol, however the relation between Vcm and power is not linear.Especially when the power in one output is relatively low, transfer fromVcm to power becomes lower and becomes even zero when power in oneoutput goes to zero. This makes it more difficult to get the desiredorthogonal control and may even make it impossible to keep both outputswell regulated when one output is at low power.

An embodiment will now be described that allows for regulating bothoutputs at low power levels by making the transfer from control inputsto power more linear for both control inputs. This embodiment mayinclude the following features: a switch mode power converter with atleast two outputs including a regulated system voltage output and asecond regulated output, where the second output can be for a highervoltage load with a regulated current; the switch mode power convertermay be controlled by state variables, Vcm, Vdm, where the Vcm variableis included in a local feedback loop in order to improve linearity ofthe transfer from control input to power; and a method of how to limitthe control variables if the local feedback loop cannot be kept closedbecause of a zero gain situation.

FIGS. 7A and 7B illustrate plots of Vcm and Vdm versus the powersupplied to an LED (Pled) string for various LED voltage levels. It isnoted that the LED application is just an example used herein and thatother applications are possible. More specifically, FIGS. 7A and 7Billustrate a measurement result of a practical dual output resonantconverter with a first output having a fixed load at 13V and a secondoutput driving a variable current LED string. Plots are shown for threedifferent LED strings with forward voltage of 80V, 90V and 100V for agiven nominal LED current of 200 mA and Vsupply=Vbus=380V. As shown inFIG. 7B, Vdm varies almost linearly versus Pled. On the other hand, asshown in FIG. 7A, Vcm has a nonlinear relationship to Pled, and thus howthe power is divided between both outputs has a nonlinear relationshipversus Pled.

This nonlinear relationship leads to values for k12 and k22 of thecontroller matrix that depend on the operating point of the converter.Because the values for k11, k12, k21, and k22 are based upon thederivatives of the Vcm and Vdm curves, if the curve is linear, thenthese values remain constant throughout the range of operation. If thecurve is nonlinear, then the values of k11, k12, k21, and k22 will varydepending on the specific operating point. This makes it difficult oreven impossible in practice to maintain the orthogonal control that isrequired to prevent load steps in output of the dual output resonantconverter. For example, load steps in the output voltages of the dualoutput resonant converter for an LED TV result in visible lightvariations in the LCD backlight or disturbance in the 12V supply asresult of PWM regulation of the LED strings.

FIG. 8 illustrates embodiment of a dual output controller for a dualoutput resonant converter that linearizes Vcm. The dual outputcontroller 800 is an extension of the dual output controller 220 of FIG.2. The dual output controller 800 includes an error processor 805 thatoperates like the dual output controller 220 of FIG. 2. The errorprocessor 805 receives error signals that tracks the output of the dualoutput resonant converter as described above. The error processor 805computes the value of Vdm as described above. The error processor 805also computes a value Δ Ec_desired as ΔEc_desired=error1·k21+error2·k22. Δ Ec_desired is related to the desireddifference between the two output powers of the dual outputs of the dualoutput resonant converter. This power difference may be based upon theoutput voltages, output currents, or difference in converted energyduring each half-cycle; accordingly, Δ Ec_desired may be based upon anyof these measures.

The dual output controller 800 also includes a local feedback loop 810that adapts Vcm such that the variable related to the difference inoutput power based upon Δ Ec_desired. The local feedback loop includesan adder 815 that takes the difference between Δ Ec_desired and themeasured difference between the two output powers of the dual outputs.This difference is then fed into a regulator 820 that produces a valuefor Vcm that is then used by the dual output resonant converter asdescribed above. The regulator 810 is shown as a proportionalintegration (PI) regulator, but other types of regulators may be used aswell, such as for example, proportional integrators, differentiationregulators, etc.

This feedback loop results in Δ Ec_desired having linear relationshipwith respect to Pled as long as the loop is closed. This results in anew dual output controller 800 with inputs Vdm and

Ec_desired resulting in orthogonal control for the total system forevery operating point without the need to vary k11, k21, k12, k22 basedupon the specific operating point.

Now it will be shown that this feedback loop results in a linearrelationship between Δ Ec_desired and Pled as desired. First assume that

Ec_desired is expressed as the difference in power between the twooutputs, then assuming the 13V output is loaded with 13V, 2 A=26 watts,then

Ec_desired=0, means power in the LED output also equals 26 watts. Alsousing the curve for an led voltage of 90V, then Pled=0 corresponds toVcm=215V, which is also the point above which Vcm can no longer definethe power, because all power flows in the 13V output. This leads to anew curve defining the relationship between

Ec_desired and Vcm as result of the feedback loop 810. FIG. 9illustrates the relationship between

Ec_desired and Vcm as result of the feedback loop. FIG. 9 indicates thevertical asymptotic behavior for Vcm=f(

Ec_desired) when led power goes to zero. This then leads to FIG. 10which illustrates the linear relationship between

Ec_desired and Pled. This linear relationship means that Vcm will havethe same original nonlinear relationship with Pled, because the pathfrom Vcm to the resonant controller did not change by adding the localfeedback, however the feedback loop 810 drives the error signal that isthe input for the PI regulator 820, therefore the regulator adapts Vcmsuch that the desired behavior of curve below occurs until

Ec_desired and

Ec=poutLED−Pout13V) are equal and the error signal goes to 0.

In a dual output resonant converter that used duty cycle and frequencycontrol, the output powers have a nonlinear relationship to the dutycycle and frequency control parameters. As a result, the feedback loop810 of FIG. 8 may also be applied to the duty cycle and frequencycontrol parameters. FIG. 11 illustrates dual output resonant converter1100 the using feedback loops to linearize the duty cycle and frequencycontrol parameters. The dual resonant converter 1100 includes afrequency and duty cycle controlled core 1105, PI regulators 1110 and1115, and adders 1120 and 1125. The first feedback loop includes adder1120 and PI regulator 1110 and receives an input of the total powerdesired and a measure of the total power produced. The first feedbackloop drives the frequency control parameter to a value that leads to thetotal power desired. Likewise, the second feedback loop includes adder1125 and PI regulator 1120 and receives an input of the difference inpower desired and a measure of the difference in power produced. Thesecond feedback loop drives the duty cycle control parameter to a valuethat leads to the difference in output desired. As a result, the two newcontrol inputs Power_desired and

Ec_desired now have a linear relationship with the output powers whilethe nonlinear relation between the original control parametersfrequency, duty cycle are still nonlinear with power.

The curves of FIGS. 9 and 10 further show that at the edges where one ofthe output powers go to zero, the feedback loop 810 cannot be keptclosed anymore, because even for an infinite amount of variation of Vcm,zero variation of

Ec still occurs, i.e., vertical asymptotic behavior. From FIG. 7A it canbe seen that for values of Vcm above a certain level the gain from Vcmto Pout1-Pout2 becomes zero. At that situation all power flows to oneoutput. This also means that the feedback loop 810 cannot be keptclosed, because a change in Vcm will not have any effect anymore on theoutput currents. The result will be that Pout1 or Pout2 (depending onwhich of the two is zero) is only controlled by the Vdm signal; howeverwithout additional measures, Vcm will continue to adapt while entering adead zone.

As it is possible to control one output by one control signal, thissituation can be used in practice when it is possible to keep the openfeedback loop in a well-defined state as close as possible to the pointwhere the influence of Vcm on output power is lost. In this way rapidrecovery is possible without too much transient effects to the closedloop situation when the zero power at one output situation is left.

An embodiment will now be described where the Vcm signal is clamped at amaximum or minimum value when it is detected that the output power atone power output has become zero. In this situation using a clamped Vcm,it is possible to keep the loaded output regulated.

Clamping can for example be done at a predetermined level just outsidethe normal operating region. It is also possible to do the clampingbased on information related to the output current. The embodiments forclamping Vcm may use, for example, the following information: 1) actualsensing of the output current and limiting Vcm when the output currentgets close to zero; 2) sensing of the voltage at an auxiliary windingand detecting if the voltages gets larger than the reflected outputvoltage during a certain time interval during a half-cycle; and 3)checking if continuous conduction mode (CCM) operation occurs. This maybe accomplished by a power detector that detects when the power at oneof the dual outputs goes to zero, and then a limit detector that limitswith value of Vcm when the power detector indicates that the power atone of the dual outputs has gone to zero.

FIG. 12 illustrates a block diagram of a first embodiment for clampingVcm. In this embodiment, the output current is sensed and compared witha threshold in order to generate an error signal that overrules thestandard feedback loop, for example by taking the maximum of thestandard Vcm signal and the error signal to generate a signal Vcm_limthat is used by the dual output resonant converter instead of the normalVcm signal. The block diagram of FIG. 12 includes a feedback loop 1210the same as shown in FIG. 8 and that operates as described in FIG. 8 andthat includes a PI regulator 1215 and adder 1220. The dual outputresonant converter 1205 is also shown. A limit detector 1225 and a powerdetector 1230 are also included. The power detector 1230 is a circuitthat receives a measurement of the output power of one of the outputs ofdual output resonant converter 1205. This measurement may be current,voltage, or power depending on the specific implementation. The powerdetector 1230 is a circuit that also receives a reference value thatcorresponds to a nearly zero power output of the one output. The powerdetector 1230 produces a value indicating that the output power isnearly zero. The limit detector 1225 outputs a Vcm_lim signal. ThisVcm_lim signal corresponds to Vcm during normal operation, but when theoutput of the power detector 1230 indicates that the output power iswithin a threshold value of zero power, then the limit detector 1225outputs a Vcm_lim value that is a fixed maximum value for Vcm to thusclamp the value of Vcm_lim. The limit detector 1225 also outputs an overrule signal that is sent to the PI regulator 1215 that stops theoperation of the PI regulator 1215 until the output power increasesabove the threshold value again. In this case, further integration ofVcm and therefore entering a dead zone is prevented. Preventing such adeadzone is preferred in order to prevent long dead times when theoutput power increases above the threshold value again

FIG. 12 shows the basic block diagram for one power output only. For theother power output a similar circuit would be used, but the limitdetector would instead detect a minimum Vcm value as the Vcm term hasthe opposite effect on the power in each power output.

FIG. 13 illustrates a block diagram of a second embodiment for clampingVcm. In this embodiment, the voltage at an auxiliary winding that iscoupled to the secondary winding is monitored in order to generate anerror signal for limiting the Vcm signal. The block diagram of FIG. 13includes a feedback loop 1310 the same as shown in FIG. 8 and thatoperates as described in FIG. 8 and that includes a PI regulator 1315and adder 1320. The dual output resonant converter 1305 is also shown. Alimit detector 1325 and an auxiliary voltage detector 1330 are alsoincluded.

The auxiliary coil is wrapped around the common core and is coupled tothe secondary winding. Accordingly, a voltage and current is induced onthe auxiliary coil during operation of the dual output resonantconverter 1305. This auxiliary voltage may be monitored in order todetermine when the power at one output of the dual output resonantconverter 1305 goes to zero.

It can be shown that the voltage across the aux winding reaches a fixedvalue during an interval that the output current flows. This is thereflected output voltage that is visible at the auxiliary winding.Outside that interval, the auxiliary voltage is lower, because thesecondary diode is not conducting then. The auxiliary voltage is thenequal to the voltage across the primary side of the transformer. Sobased on this observation, the interval where no secondary current flowsis related to the auxiliary voltage during this interval being lowerthan the auxiliary voltage during the interval where output currentflows. This difference may be used to determine when there is no currentflowing to one of the outputs.

While FIGS. 12 and 13 both illustrate embodiments using a feedback loop,such a feedback loop is not necessary, and the limit detector may beapplied to systems that do not use this feedback loop.

The auxiliary voltage detector 1330 is a circuit that receives theauxiliary voltage value from the auxiliary coil and that acts as a powerdetector. FIG. 14 illustrates a block diagram of the auxiliary voltagedetector. The auxiliary voltage detector includes a low pass (LP) filter1405, a sample and hold (S&H) circuit 1410, a peak detector 1415, anadder 1420, and an error amplifier 1425. The LP filter 1405 receives theauxiliary coil voltage Vaux and filters out any high frequencycomponents from the auxiliary voltage signal. The output of the LPfilter 1405 is fed into the peak detector 1415, the sample and holdcircuit 1410, and the error amplifier 1425. The peak detector 1415monitors the value of the filtered auxiliary coil voltage Vaux to detectwhen a peak value is reached, and when such a value is reached, the peakdetector 1415 sends a sample signal to the sample and hold circuit 1410which samples the filtered auxiliary coil voltage Vaux. This is doneduring a first cycle where power is being transferred. The sampledauxiliary coil voltage Vaux_sampled is then input to the adder 1420. Theadder 1420 subtracts a delta value from the Vaux_sampled value, and theresults is fed into the error amplifier 1425. The output of the adder1420 is a value that is used to determine when the filtered auxiliarycoil voltage Vaux falls below a certain value relative to the maximumVaux value that indicates that the output power being monitored isapproaching zero. In a next cycle where the output power may approachzero. The error amplifier 1425 uses the output of the adder and thefiltered auxiliary coil voltage Vaux to generate an error signal. Due tothe sampled Vaux at the peak value with a small delta subtracted, theerror amplifier input from 1420 defines a slicing level with respect tothe other amplifier input that defines an interval during next switchingcycle where the filtered Vaux is above this slicing level for a shortinterval around the top of Vaux. The duration of this interval thereforedefines how close to zero the output current is and sets the output ofthe error amplifier. Therefore the error output is a duty cycle based onthe fact that Vaux is above or below the sampled value minus delta canbe used to limit the Vcm signal. The error signal output from the erroramplifier 1425 is therefore input to the limit detector 1325.

The limit detector 1325 outputs a Vcm_lim signal. This Vcm_lim signalcorresponds to Vcm during normal operation, but when the output of theauxiliary voltage detector 1330 indicates that the output power iswithin a threshold value of zero power, then the limit detector 1325outputs a Vcm_lim value that is a fixed maximum value for Vcm to thusclamp the value of Vcm_lim. The limit detector 1325 also outputs anoverrule signal that is sent to the PI regulator 1315 that stops theoperation of the PI regulator 1315 until the output power increasesabove the threshold value again. In this case further integration of Vcmends, therefore entering a dead zone is prevented similar to theembodiment of FIG. 12.

The more the output current reduces to zero, the smaller the intervalwhere Vaux gets larger than the sampled Vaux minus delta becomes, sotherefore, the duty cycle becomes smaller.

Therefore the duty cycle of the interval related to the half-cycle canthen be used as error signal for limiting Vcm.

Also in this embodiment relation between Vcm and sensed signal is lostat no load, because then the peak value of the aux voltage is not theoutput voltage anymore when the secondary diode does not conduct. Soalso here signal cannot directly be used to regulate to no load, becausefrom a no load situation it is not possible to detect the actualdistance to no load point as for every further adaption of Vcm into noload region, Vaux peak value is not related to Vout anymore, so it isnot possible to relate the duty cycle at the erroramp output to thisdistance.

For the other power output a similar error signal can be generated butwith opposite polarity similar to the first embodiment.

It is noted that the direct use of output current has a limitation thatno information is available about the distance of a given Vcm to thepoint of no load. Therefore, regulation all the way to zero load is notpossible.

A third embodiment of Vcm clamping using CCM monitoring will now bedescribed. This third embodiment utilizes the feature that the halfcycle with largest output current will enter CCM when the asymmetry islarger than a certain maximum.

FIG. 15 illustrates a plot of current on the primary coil andmagnetizing current component as result of magnetizing the core in thesituation when both power outputs are loaded. The secondary current isthe difference between the primary current and the current in the core.Accordingly, when the primary current and the core current are equal,the secondary current is zero. When point A where the secondary currentbecomes zero is before the end of the half-cycle B, the half-cycle is indiscontinuous mode (DCM).

FIG. 16 illustrates a plot of current on the primary coil and in thecore in the situation when one output has gone to zero output power. InFIG. 16, point A where the secondary current becomes zero is at the endof the half-cycle B, so the half-cycle is in CCM. Also the depth of CCM(i.e., how large the secondary current is at the end of the secondarystroke) increases when asymmetry (which is set by Vcm term) is furtherincreased.

When the resonant tank of the dual output resonant converter isdimensioned such that CCM operation occurs only when the output currentduring the other half-cycle has already become zero, then the pointwhere CCM occurs (and also the CCM depth) can be used to limit the Vcmterm. This dimensioning can be realized by choosing the proper reflectedoutput voltage (turns ratio) in combination with the required maximumsupply voltage of the LLC converter and minimum output voltage. It isnoted that CCM operation first occurs at maximum ratio between supplyvoltage and output voltage.

Detection of CCM operation is possible by sensing the output current atthe end of the secondary stroke. As an alternative the aux voltage at anauxiliary coil coupled to the secondary as described above may be sensedand checked to determine if the aux voltage decreases before (DCM) orafter the end of the secondary stroke (CCM). The depth of CCM is relatedto the time it takes after the end of the half-cycle before the auxvoltage reacts. By measuring the time difference between end of thehalf-cycle and moment that aux voltage starts to react, an error signalcan be created that is used to determine with the output power goes tozero and to then clamp the value of Vcm.

Simulations can be used to show the behavior of Vaux and Vhb, which isthe voltage of the switching node, relative to when CCM occurs. In oneexample of a simulation result of a dual output resonant converter witha 13V low voltage output and a 90V LED string output voltage based onthe controller of FIG. 8, the current in the Led output is reduced insteps. In this simulation, the Vaux crosses 0 V 75 ns after the Vhbslope starts. In this case during the half-cycle DCM occurs.

In another simulation situation CCM occurs with a delay of 200 nsbetween Vhb slope starts and Vaux crosses 0V. Another simulationsituation illustrates the situation where CCM occurs with significantoutput current at the end of the secondary stroke resulting in a delayof 265 ns between Vhb slope starts and Vaux crosses 0V. Thesecharacteristics may be used to determine when the output power goes tozero.

Using this CCM detection method of determining time differences betweenwhen Vaux crosses 0V and when a slope in Vhb starts allows for theregulation of the system to no load because at no load, the relationbetween Vcm and time difference is still defined.

FIG. 17 illustrates a block diagram of a third embodiment for clampingVcm. In this embodiment, the voltage at an auxiliary winding beingcoupled to the secondary winding and the voltage of the switching nodeVhb (see 11 in FIG. 1), are monitored in order to generate an errorsignal for limiting the Vcm signal. The block diagram of FIG. 17includes a feedback loop 1710 the same as shown in FIG. 8 and thatoperates as described in FIG. 8 and that includes a PI regulator 1715and adder 1720. The dual output resonant converter 1705 is also shown. Alimit detector 1725 and a CCM detector 1730 are also included.

The CCM detector 1730 is a circuit that receives the auxiliary voltagevalue Vaux from the auxiliary coil and the voltage at the switching nodeVhb and that acts as a power detector. FIG. 18 illustrates a blockdiagram of the CCM detector. The CCM detector 1730 includes a zerocrossing detector 1805, a start slope detector 1810, a time differencedetector 1815, and an adder 1820. The zero crossing detector 1805monitors the Vaux input voltage and outputs a time value when the Vauxinput voltage crosses 0V. The start slope detector 1810 monitors the Vhbinput voltage and outputs a time value when the Vhb voltages starts tohave a slope. The time difference detector 1815 computes a timedifference between the when Vaux crosses 0V and Vhb starts to exhibit aslope. The adder 1820 subtracts a reference value from the computed timedifference from the time difference detector 1815 to produce an errorsignal. The reference value is based upon an expected time differencethat shows that CCM operation has started. The error signal output fromthe CCM detector 1730 is then input to the limit detector 1725.

The limit detector 1725 outputs a Vcm_lim signal. This Vcm_lim signalcorresponds to Vcm during normal operation, but when the output of theCCM detector 1730 indicates that the output power is within a thresholdvalue of zero power, then the limit detector 1725 outputs a Vcm_limvalue that is a fixed maximum value for Vcm to thus clamp the value ofVcm_lim. The limit detector 1725 also outputs an over rule signal thatis sent to the PI regulator 1715 that stops the operation of the PIregulator 1715 until the output power increases above the thresholdvalue again. In this case further integration of Vcm ends, thereforeentering a dead zone is prevented similar to the embodiment of FIG. 12.

As described above, the three clamping embodiments may also be used witha converter that is controlled by duty cycle and frequency according toFIG. 11. In this case the duty cycle mainly determines how the power isdivided between the two outputs similar to the Vcm term.

In certain instances it may be difficult to control a dual outputresonant converter to ensure a full output power range for each of theoutputs. FIG. 19 illustrates a simulation result of a dual outputresonant converter being controlled by Vdm and Vcm signals as describedabove. The output voltages are such that at Vcm=0, equal powers occur atboth outputs. For simplicity, the turns ratio for both outputs are thesame such that both output currents have the same weight factor withrespect to output power.

From FIG. 19 it can be seen that for Vcm=0 both output powers are equaland proportional to Vdm. For Vcm>0, Iout2 is increased while Iout1 isreduced, basically shifting power from one output to the otherespecially for relatively small powers, for example, Vdm<approx. 0.3.This process of shifting power stops rather abruptly when Iout1 becomesclose to 0, while at larger values for Vdm, the process of shiftingpower is smoother. For Vdm>0.6 in this example, undesired effects occur,such as the switching frequency starting to change significantly andother effects that require detailed understanding of the resonant tankbehavior due to large resonance and therefore is not discussed here.Therefore, the dual output resonant converter may therefore be designedto operate preferably at Vdm below a certain maximum of 0.3-0.6referring to the scaling factor used in FIG. 19. In this operatingrange, the behavior of the dual output resonant converter is such thatsymmetry is mainly set by Vcm, while total power is mainly set by Vdm.Based on a simplified model that assumes an abrupt change over of powerand certain dimensioning of the resonant tank, power as function of bothVcm and Vdm in each output can be modelled as given in FIG. 20. FIG. 20illustrates a simplified model of the plot of FIG. 19. Although thismodel is only a very simple approximation of the actual behavior, it canbe used to get a better understanding of the main problems of providinga full range of power control for each output and how to overcome thoseproblems.

As described above, the required controller should have the inversebehavior of the converter, which can in general be written as:Vcm=error1·k21+error2·k22; andVdm=error1·k11+error2·k12.

These equations are small signal equations, so a change in error signalscauses a change in Vdm and Vcm. For a practical controller being drivenby a current from two opto-couplers, the maximum power occurs when bothopto-coupler currents are zero. In this case, the maximum value of Vdm(Vdm_max) occurs. FIG. 21 illustrates a block diagram of a dual outputcontroller showing that when the measured error signals are zero so thata value of Vdm_max is output from the controller. As the error signalsgrow, the value of Vdm decreases from Vdm_max because the power is lessthan the maximum value.

For the specific case where symmetrical behavior occurs for both outputs(i.e., the same reflected output voltage, same coupling to primary forboth secondary windings, and both outputs operating at about the sameoutput power), symmetry or the difference between the output powers ismainly set by Vcm, while the total power is mainly set by Vdm. Thismeans that that k11 and k12 are equal, while k21 and k22 have the samemagnitude but opposite signs. This situation is valid for both outputpowers larger than zero and small to moderate power levels.

For the region where Vcm causes one of the output powers to go to zero,a controller according to the equations for Vcm and Vdm above with k11and k12 equal and k21 and k22 having the same magnitude but oppositesigns can lead to problems, because both error signals can reduce theVdm signal setting the total power, while Vcm is no longer capable ofsetting the division of power between both outputs. FIG. 22A illustratesthe situation where Vcm is no longer capable of setting the division ofpower between both outputs.

Pout1 is the power in the first output and Pout2 is the power in thesecond output. In region 1 2205, power flows to both outputs and as thecontroller is set to the optimum parameters k11=k12 and k21=−k22, andPout1 is set by error signal error1, while error1 does not have aneffect on Pout2. The reason that Pout2 can be kept constant is thaterror1 reduces Vdm while error1 also adapts Vcm in the proper ratio suchthat the power reduction of Pout2 due to Vdm's reduction is compensatedby shifting power from Pout1 to Pout2 by changing the Vcm term.

In region 2 2210, however, Vcm loses its influence as Pout1 reaches 0.The result is that only the Vdm term can change the power, so the twooutput powers are then set by only one variable. Due to k11=k12 theerror1 signal still continues to reduce the total power by reducing theVdm term, while the compensating effect of k21 in combination with Vcm,shifting power from Pout1 to Pout2 is lost, as there is no power left toshift from the first output to the second output as Pout1 is alreadyzero. The result is that orthogonal control is lost because Pout2 isalso reduced by error1, which is undesired.

An embodiment that overcomes this problem of power loss in one outputand the loss of orthogonal control by limiting the reduction of the Vdmterm by each error signal to a value of, for example, half of themaximum value, such that power in the other output can be kept undercontrol by the other error signal is described below.

As before, an embodiment of a dual mode resonant converter with at leasttwo outputs including a regulated system voltage output and a secondregulated output, where the second output can be a LED string withregulated current will be used as an example. The dual mode resonantconverter is controlled by state variables Vcm and Vdm, where the Vdmvariable is limited to a minimum value by each error signal such thatthe other error signal is capable of increasing the power to therequired value. In order to get a better understanding of the details ofthis embodiment, first a more detailed analysis of the problem isdescribed.

FIG. 22B illustrates an example of the dual output resonant converterbeing controlled by Vdm and Vcm. In this example scaling factors of theconverter are set such that Vdm=100V gives total power of 200 watts (100watts in each output at symmetrical operation at Vcm=0), and a Vcm of100V shifts 100 watts of power from Pout1 to Pout2.

Now assuming a transient situation, where the load is changed from 100watts in both outputs (Vdm=100V, Vcm=0V) to no load at the Pout1 outputand 100 watts at the Pout2 output (Vdm=50V, Vcm=50V), no load at Pout1will cause the regulation loop to increase the error1 signal, howeverdue to overshoots, it is likely that also the error signal overshootsand therefore reduces Vdm below 50 by factor k11, while Vdm=50V isneeded to make Pout2=100 watts. Especially when this is a load step tono load, the error signal can over react because of overshooting of theoutput voltage, while the converter cannot produce negative power toreduce the output voltage. The error2 signal can partly compensate thepower requirement for the other channel by reducing its error signal,but as the opto-coupler current cannot become negative, measures must betaken to make at least Vdm=50V via the path of error1.

An embodiment of a dual output resonant converter will now be describedwhere the contribution of each error signal to the Vdm signal is limitedsuch that the Vdm signal cannot be made lower than required by themaximum power level in the other channel. For example taking theconverter of FIG. 22B, with the required power range for each outputbetween 0 and 100 watts (200 watts max giving Vdm_max=100V), a limit maybe set to maintain at least half of Vdm_max 50V for the error1 path toensure a power of 100 watts can be delivered by the error2 path.

Also a limit should be set to maintain at least half of Vdm_max 50V forthe error2 path to ensure a power of 100 watts can be delivered by theerror1 path. Including this limit and the fact that the Vcm term canfully shift the available power between both outputs guarantees thatboth power ranges, even when the error signals overshoot. Some marginhowever may be added to not use the full amplitude of the available Vcmterm. There are two reasons for adding this margin. First, due to thenonlinearity of the Vcm term when power in one channel comes close tozero, the margin provides a benefit. Second, the fact that output powerslightly reduces when the asymmetry increases, slightly smaller powerbecomes available than expected, so the margin can compensate for thisreduction.

Therefore Vdm_max based on total power at symmetrical operation shouldbe multiplied with an additional factor k slightly larger than 1 with,for example, a typical range of 1.1-1.2, to take the second effects intoaccount. For the first effect it is required to make the limit to thepower based on Vdm term slightly lower than required by an additionalfactor m slightly larger than 1 with, for example, a typical range of1.1-1.2, such that the error signal does not need the complete amplitudeof the Vcm term to get the power to zero by shifting the residual powerto the other output.

FIG. 23 illustrates an embodiment of a dual output controller thatlimits the value of Vdm. The dual output controller 2300 includesmultipliers 2305, 2310, 2315, and 2320, maximum detectors 2330 and 2335,Vdm limit blocks 2325 and 2340, adders, 2345, 2350, and 2360, and scaledVdm_max block 2355. The dual output controller 2300 receives the errorsignals error1 and error2. The multipliers 2305 and 2315 receive theerror1 signal as an input and multiplies the error1 signal by k11 andk21 respectively. The multipliers 2310 and 2320 receive the error2signal as an input and multiplies the error2 signal by k12 and k22respectively. The adder 2350 adds the outputs of the multipliers 2315and 2320 to produce the control signal Vcm. The maximum detectors 2330and 2335 receive values, Vdm_max*alpha/m and Vdm_max*(1−alpha)/m, fromthe limit blocks 2325 and 2340 respectively. The purpose of the factor mis described above. The maximum detectors 2330 and 2335 also receive theoutputs from multipliers 2305 and 2310 respectively. The maximumdetectors 2330 and 2335 output the input maximum value of the inputsreceived. The adder 2345 then adds the outputs of the maximum detectors2330 and 2335. Then, adder 2360 subtracts the output from the adder 2345from the value from the scaled Vdm_max block (Vdm_max*k). The purpose ofthe factor k is explained above.

If a value of 0.5 is selected for alpha, each output may produce a powerbetween zero and half of the maximum power. If larger range of power isrequired for one output, a value for alpha different from 0.5 may bechosen. For example, taking a converter with the characteristics shownin FIG. 22 having a power range for the first output (corresponding toerror1) between 0 and 150 watts and for the second output (correspondingto error2) between 0 and 50 watts (for a 200 watts max givingVdm_max=100), a limit should be set for the error1 path to reserve atleast 50 watts of power (Vdm_max=25) for the error2 path to ensure apower of 50 watts can be delivered by the error2 path when error2 iszero. Also a limit should be set on the error2 path to reserve at least150 watts of power (Vdm_max=75) for the error1 path to ensure a power of150 watts can be delivered by the error1 path when error1 is zero. Thispower range can be set by the factor alpha between 0 and 1.

The embodiment of FIG. 23 may be combined in various combinations withthe embodiments of FIGS. 8, 12, 13, and 17, where the dual outputcontroller 2300 replaces the dual output controllers in thoseembodiments, to thus add the features of limiting power associated witheach output with the features of linearizing and clamping the controlvariable Vcm. In such a case, the output of the adder 2350 becomes thevalue

Ec_desired used in the feedback loop.

Further, the embodiment of FIG. 23 may be combined in variouscombinations with the embodiments of FIGS. 11, 12, 13, and 17, where thedual output controller 2300 replaces the dual output controllers inthose embodiments, to thus add the features of limiting power associatedwith each output with the features of linearizing and clamping thecontrol variables duty cycle and frequency. In such a case, the outputof the adder 2350 becomes the

Ec_desired value used in the feedback loop, and the output of adder 2360becomes the total power value used in the feedback loop.

FIGS. 6B and 6C illustrate dual output resonant converters using Vcapcontrol and Vcr control. Both methods are used in practical resonantconverters and are basically compatible, although there are slightdifferences, for example with the Vcr method, the DC component of Vcr isequal to Vsupply/2×duty cycle of the switching node, while the DCcomponent using the Vcap method is zero (because it is a voltage acrossan inductor). As a capacitive divider is often used to reduce the signalamplitude of a few 100 volts to levels of a few volts as required forfurther processing, it could be an advantage to use the Vcap method asthen the DC component does not have to be reconstructed.

For proper control of the Vcm control variable, the proper DC levelneeds to be applied. In a symmetrical situation with equal power at bothoutputs, Vcm equals 0. In practical situations however there are severalreasons why the operation of the dual output resonant converter may notbe symmetrical. First, there can be asymmetry in the transformer,because the physical windings of the secondary side cannot be at thesame location. Second, the output voltages in a dual output resonantconverter can be different from the symmetrical situation with equalreflected output voltages. Third, when the resonant capacitor is placedat ground side of the transformer, the DC value of the voltage acrossthe resonant capacitor equals Vsupply/2×duty cycle, where duty cycle isthe duration of a half-cycle related to the total period. Often acapacitive divider is used to divide the voltage to a low value that canbe processed by the controller IC. Such a capacitive divider cannotdivide the DC component, so therefore also the DC component of Vcm isundefined or badly defined. A resistive divider can be placed inparallel to define the DC component, however mismatch between bothfactors of the two voltage dividers due to component tolerances stillresults in too much asymmetry. Further, the resistive divider mayintroduce a phase shift due to parasitic capacitances or consume toomuch power because of its high resistance.

In a normal resonant converter with symmetrical operation, the DCcomponent may be reconstructed by connecting the capacitive divider to afixed voltage with a high resistance resistor and then adding anadaptive DC component based on measuring the duty cycle or ratio ofcurrents in both half-cycles. For a dual output resonant converter,different powers occur for each half-cycle, so this method ofreconstruction cannot be used.

An embodiment of a dual output resonant converter will now be describedthat reconstructs the proper DC component such that the required powercan be delivered to both outputs. Features of this embodiment include: acheck if one of the error signals is clipping to a minimum or maximumvalue as an indication that the desired power cannot be delivered; acheck of a signal related to the each output indicating that an outputreaches a no load situation; adding an additional offset term to the Vcmsignal or to values that Vcm is compared to; a way to adapt theadditional offset depending on clipping of one of the error signals; andthe direction in which the offset is adapted is set by both clipping ofthe error signals and at what level in combination with informationabout each output reaching a no load situation or not.

FIG. 24 illustrates a first embodiment of a dual output controller usinga Vcm offset. This dual output controller 2400 is similar to other dualoutput controllers described above that implement the calculations forVdm and Vcm based upon the error1 and error2 signals. Multipliers 2405,2410, 2415, and 2420 variously multiply the error1 signals and error2signals by constants k11, k12, k21, and k22 as shown. The adder 2425adds outputs of multipliers 2405 and 2410 together to produce Vdm. Theadder 2430 subtracts the outputs of multipliers 2415 and 2420 from oneanother to produce an initial value of Vcm. The adder 2435 then adds aVcm_offset value 2440 to the output of the adder 2430 to produce Vcm.

When the DC component of the Vcm term is wrong too much power may go toone output, while too little power goes to the other output as opposedto the desired output values for the two outputs. For that reason, theadditional term Vcm_offset is added to the Vcm control variable in orderto shift power from one output to the other if such a mismatchedsituation occurs.

FIG. 25 illustrates a second embodiment of a dual output controllerusing a Vcm offset. This dual output controller 2500 is similar to thedual output controllers described above in FIG. 24, but insteadimplements the Vcm_offset is implemented in the dual output resonantconverter. The dual output controller 2500 produces values of Vdm andVcm that implement the calculations for Vdm and Vcm based upon theerror1 and error2 signals as described above. In this second embodiment,the Vcm_offset is used later in controlling the switching of theswitches in the dual output resonant controller. This addition of theVcm_offset may be implemented by injecting a current or charge into thecapacitive divider described above used to measure Vcr. Morespecifically, this may be accomplished by the combination of thecombination block 2505, comparators 2510 and 2515, adder 2520,Vcm_offset generator 2440, and the driver control logic 2525.

FIG. 26 illustrates an embodiment of a Vcm_offset circuit. TheVcm_offset circuit 2600 includes comparators 2605, 2610, 2615, and 2620,OR gates 2625 and 2630, AND gates 2635 and 2640, and integrator 2645.Comparator 2605 receives the error1 signal and an error1 minimum valueto produce an output signal A that indicates that the maximum outputpower that can be delivered at output 1 is too low. Comparator 2610receives the error1 signal and an error1 maximum value to produce anoutput signal B that indicates that the minimum output power that can bedelivered at output 1 is too high. Comparator 2615 receives the error2signal and an error2 minimum value to produce an output signal C thatindicates that the maximum output power that can be delivered at output2 is too low. Comparator 2620 receives the error2 signal and an error2maximum value to produce an output signal D that indicates that theminimum output power that can be delivered at output 2 is too high.Under normal conditions, both error signals error1 and error 2 have avalue between a minimum value and a maximum value. A regulated outputVout1 for example causes an error signal error1 to be within theminimum-maximum range as long as the load connected to Vout1 matches thepower delivered by the converter to that output. When the load isincreased above the maximum power that the output can deliver, the errorsignal adapts to the minimum or maximum, depending on the chosenpolarity of the error signal. It is assumed, for example, that theerror1 signal being at the maximum causes the maximum power to bedelivered and the error1 signal being at the minimum causes the minimumpower to be delivered. A properly dimensioned converter is capable ofdelivering the required maximum and minimum power, so then an errorsignal reaching maximum or minimum should not occur during steady statesituations and means that the offset level is not correct. Duringtransients it is possible that for a short interval the minimum ormaximum levels are crossed, but depending on the chosen bandwidth of themain loop, this situation cannot take longer than a certain time, forexample 500 usec. Only during startup or after a specific faultcondition could it could take longer.

So if it is detected that an error signal is outside the limits for acertain time longer than an expected time, there could be a problem withthe Vcm term and then the action is to slowly adapt the Vcm_offsetsignal in the required direction in order to shift power from one outputto the other. As the main regulation loop also controls power deliveryto both outputs, it is required that the Vcm_offset shifting mechanismhas a bandwidth significantly lower than the bandwidth of the main loopto prevent instability due to interaction of both loops. OR gate 2625receives the outputs A and D and counts if either A or D or A and Dindicate that error1 and/or error 2 are out of range. If so, the OR gate2625 produces an output that causes the value of the Vcm_offset term tochange. OR gate 2630 receives the outputs B and C and counts if either Bor C or B and C indicate that error1 and/or error 2 are out of range. Ifso, the OR gate 2630 produces an output that causes the value of theVcm_offset term to change.

The AND gate 3635 receives the output of the OR gate 2625 and anindication that the output current at the second output Iout2 is greaterthan zero and produces an AND of the inputs. The AND gate 3640 receivesthe output of the OR gate 2630 and an indication that the output currentat the first output Iout1 is greater than zero and produces an AND ofthe inputs.

Integrator 2745 receives the outputs of the AND gates 3635 and 3640 andproduces a value of Vcm_offset as an output. The integrator 2645 may beimplemented as an up/down counter. As the AND gates 2635 and 2640produce output values, the integrator 2645 adjusts the value ofVcm_offset to compensate for the errors in the output powers. The logicfunction of FIG. 26 assumes a certain polarity of the error signalserror1 and error2, so with a different polarity a different logiccombination occurs, but the idea remains the same. The signal Vcm_offsetmay be for example a voltage, current, charge, or a digital value. Otherimplementations of the integrator 2645 may be used as well.

The signals Iout1>0 and Iout2>0 are included to prevent the activationof the Vcm_offset adaption as a zero load occurs, because in suchsituations the regulation loop always reacts in this way so when Iout=0has been reached, the corresponding shift action is prevented by thelogic, because it was already clear that the Vcm_offset term was capableof making no load.

In another embodiment, an additional functionality may be added when thesystem cannot deliver sufficient power to both channels. First, theaction may be to do nothing. Second the values of k11 and k12 may beincreased.

The embodiment of FIG. 26 using the Vcm_offset term may also be combinedin various combinations with the embodiments of FIGS. 8, 12, 13, and 17.Further, the embodiment of FIG. 26 may be combined in variouscombinations with the embodiments of FIGS. 11, 12, 13, and 17. In thisembodiment a signal duty cycle offset may be added so the duty cycle inFIG. 11.

In the embodiment of FIG. 8 a dual output resonant converter with alocal feedback loop is presented to improve the orthogonal controlperformance as needed to prevent cross regulation problems between bothoutputs. At a no load situation for one of the two outputs, this newlocal feedback loop has zero loop gain, so the local feedback loop thenbecomes an open loop, causing one of the resonant converter controlvariables (Vcm) to drift away resulting in undesired effects, such as atime interval where the output power cannot be controlled properly,because Vcm change has no influence on output power, thereby disturbingthe regulation loop for both outputs. In the embodiment of FIG. 2. 24-26several solutions are proposed to limit the Vcm term such that theconverter can deal with a no load situation at one output. Although bothnormal operation and no load operation at one output are significantlyimproved by the embodiment of FIG. 8, one problem remaining is that thechangeover between both situations does give undesired transienteffects. Such transient effects may disturb both outputs, for example,both the LED and low voltage outputs when used in an LCD TV with PWMdimming.

Now an embodiment that solves this problem in a more fundamental way bylimiting the corresponding error signal instead of the Vcm signal willnow be described. This embodiment leads to a much better performanceduring the changeover situation between normal operation and no load atone output, because the required relation between Vcm and Vdm terms ismaintained also during the changeover.

This embodiment may include the following features. The dual outputresonant converter is controlled by state variables Vcm and Vdm, wherethe Vcm variable is included in a local feedback loop in order toimprove linearity of the transfer from control input to power asdiscussed above with respect to FIG. 8. The embodiment includes adetector that detects if one of the outputs goes to a no load situation.Also, the embodiment includes a processing block including a samplingfunction that samples the Vcm signal when no load situation is reachedor almost reached. The embodiment includes a local regulation loop thatuses the sampled Vcm signal as reference and regulates/limits the actualVcm value to a value slightly above the reference. This regulation loopgenerates a second error signal that is compared with the error signalfrom the main regulation loop. The minimum of both error signals is thentaken as input for the controller such that the controller generates theproper Vcm and Vdm signal and the proper relation in between them tomaintain orthogonal control.

The embodiment of FIG. 8 using local feedback to control Vcm is used asthe basis for this embodiment. FIG. 27 illustrates an embodiment of aconverter that limits the error signals. The converter 2700 includes adual output resonant converter 2720, an error processor 805, and afeedback loop 810 including an adder 820 and regulator 815 as describedabove with respect to FIG. 8. The converter 2700 further includes zeropower detector 2705 and a zero output power processor 2710. The zeropower detector 2705 detects when the power at either of the two outputsof the dual output resonant converter 2720 becomes zero and outputs asignal indicating this condition. The zero power processor 2710 receivesthe two error signals error1 and error2 and produces limited errorsignals error1_lim and error2_lim based upon the output signal from thezero power detector 2705.

FIG. 28 illustrates an embodiment of zero output power processor. Thezero output power processor 2800 includes a track and hold circuit 2805,an adder 2810, a regulator 2815, and minimum detection circuit 2820. Thezero output power processor 2800 receives an error signal that may beeither the error1 signal or the error2 signal. Processing for only oneerror signal is shown, but the same processing would be performed on theother error signal in the same manner. The error signals error1 anderror2 indicate the status of the outputs at the secondary side of thedual output resonant converter.

The track and hold circuit 2805 receives the current value of Vcm as aninput along with the output of the zero output detector and outputs avalue Vcmreg. When the output power becomes zero, then the track andhold circuit 2805 sets the value of Vcmreg to the value of Vcm at thetime the output power become zero. Otherwise, the track and hold circuit2805 simply sets the value of Vcmreg to the current value of Vcm in atracking mode. The adder 2810 computes the value Vcmreg+Vcmoffset−Vcm.The output of the adder 2810 is input to the regulator 2815 which thenproduces a signal indicative of the error signal. The regulator may beany type of regulator including, for example, PI regulators,proportional integrators, differentiation regulators, etc. The minimumdetector 2820 receives the error signal and the output of the regulator2815 and outputs the minimum value of the two inputs.

As long as power is delivered to both outputs of the dual outputresonant converter 2720, the track and hold circuit 2805 is in trackmode, which means that the signal Vcmreg equals Vcm and the adder outputequals the Vcmoffset signal which is the input for regulator 2815. Theoutput of 2815 then integrates to a high level that is always largerthan the error signal such that the error signal directly passes to thelimited error signal. As optional feature, the Vcmreg output is onlyupdated at a certain time during the switching cycle as Vcm includesripple due to the main regulation loop reacting to the output currentpulses of the converter. The minimum detector 2820 input from theregulator 2815 is then high and the error signal is directly passed tothe limited error signal output. In this situation, the zero powerdetector 2705 does not affect the operation of the dual output resonantconverter 2720. In this embodiment, a minimum detector is used relatedto the polarity of the error signal. It is noted that the oppositepolarity of the error signal is also possible and then a maximumdetector would be used instead of the minimum detector.

When one output of the dual output resonant converter 2720 comes closeto zero load, this is detected by the zero power detector 2705. Thisdetected event is then processed by the zero output power processor 2710as follows. First the track and hold circuit 2805 holds value of the Vcmterm in the value Vcmreg. Now the difference between Vcm and Vcmregdrives the regulator 2815 and overrules the normal error signal as longas Vcm is on the wrong side of Vcmreg+Vcmoffset. The result is that thislocal feedback loop sets the limited error signal to the proper valuesuch that the proper Vcm signal occurs when just getting to no load. Dueto the Vcmoffset term, Vcm will settle at a value just below the borderof zero power. As the local loop allows both positive and negativevalues of Vcmreg−Vcm+Vcmoffset, it is possible to limit the systemreally at zero output power with some margin.

The zero power detector 2705 may be implemented as described in FIGS.13, 14, 17, and 18, which include: the actual sensing of the outputcurrent and limit Vcm when output current gets close to zero; sensing ofthe voltage at an aux winding and detect if the voltages gets largerthan the reflected output voltage during a certain time interval duringa half-cycle; or checking if CCM operation occurs.

Although the zero output power processor 2710 according to FIG. 28 canactually regulate the output current to zero, the relationship betweenVcm and output current can change due to for example changing supplyvoltage of the converter or changing output voltage, because of a loadconnected to the output where the output voltage is temperature andcurrent dependent. Therefore it is not sufficient to sample Vcm onlyonce when zero load occurs.

This problem may be solved by regularly driving the loop towards thepoint where power starts to flow and then resampling the correspondingVcm value. There are several options possible to realize this, forexample by overruling the error signal of slowly adapting the level ofVcmoffset.

One embodiment based on slowly adapting the level of Vcmoffset is givenin FIG. 29. Based on zero output power processor 2710 as shown in FIG.28, the following items are added: a ramp generator 2905 and adders2910, 2915, and 2920. The track and hold circuit 2805, regulator 2815,and the minimum detector 2820 are the same as described with respect toFIG. 28.

An additional input resample is added that drives the ramp generator2905. This ramp generator 2905 generates an additional offset added toVcmoffset by adder 2910 making a total offset Vcmoffset1. As the idea isto go back to the border where power is just delivered, the rampgenerator 2905 produces a negative ramp effectively compensatingVcmoffset.

When the resample pulse is received, the ramp generator 2905 generates aramp signal that starts from zero, slowly increasing its amplitude. Theresult will be that the regulator starts adapting its output such thatthe limited error signal changes such that the controller startsadapting Vcm in the direction of increasing the power at the no loadoutput. As a result, the Iout=0 signal will change to zero when the rampreaches a sufficient amplitude, indicating that power starts to flow inthe output.

When Iout=0 goes false (output current starts to flow) then the trackand hold circuit 2805 will enter track mode and therefore starts passingthe actual value of Vcm to its output Vcmreg as long as Iout=0 is false.

At the same time as result of Iout=0 going false, the ramp is stoppedand reset to zero. So then the original Vcmoffset is added to Vcmreg.This offset being integrated by the regulator 2815 overrules the errorsignal again and reduces the power at the output until finally itreaches zero again. As result, Iout=0 changes to true and causes thetrack and hold circuit 2805 to hold the last Vcm value that now againcorresponds to the output power being equal to zero. Including theVcmoffset term guarantees that the local loop including regulator 2815and minimum detector 2820 causes Vcm to adapt to a level giving no loadagain.

This procedure is repeated regularly. As power delivery can be detectedwithin a few switching cycles while the power produced can be very lowdue to a limited ramp-up speed, the effective power generated can belimited to less than 50 mwatt for a 100 watt led power system This smallpower can easily be dissipated in a parallel resistor or additional loadconnected to a second winding with a rectified output generating powerduring the same half-cycle as the LED output, so the LEDs are kept fullyoff.

The procedure can also be activated when an event is detected from whichit can be expected that the relation between Vcm and the point of nopower delivery has changed, for example after a mains dip or a largeload step.

If for whatever reason power starts to flow in the output that wasregulated to no load, this will automatically cause the steps of Iout=0going false to Iout=0 going back to true to happen thereby automaticallyupdating to the proper Vcm and Vcmreg level.

The embodiment of FIG. 27 may also be applied more generally such as forexample with a dual output resonant converter core being controlled byduty cycle and switching frequency such as illustrated in FIG. 11. FIG.30 illustrates an embodiment of a converter that limits the errorsignals using duty cycle and frequency controls. Such a converter may becontrolled by a similar controller with one or more local feedback loopsin order to linearize the transfer from control input to outputvariable, such as given in FIG. 30 including two feedback loops for bothfrequency and duty cycle. The converter in FIG. 30 operates the same asthe converter in FIGS. 27-29. In this case the duty cycle variabledrives the asymmetry of the converter instead of Vcm and thereforerelates to the point where one output goes to no load. In this case theduty cycle can then be used as input for the input of the track and holdcircuit 2805.

Although the outputs of the resonant converter are described in terms ofoutput voltages, Vout1 and Vout2, it should be understood that thecontrol techniques described herein are applicable to and may beimplemented in response to the voltage at the outputs, Vout1 and Vout2,the currents at the outputs, Iout1 and Iout2, the power at outputs,Pout1 and Pout2, or some combination thereof. In some instances, theterm “voltage/current” is used to refer to the voltage and/or thecurrent, such that the term may refer to the voltage, may refer to thecurrent, or may refer to both the voltage and the current. The term mayalso refer to power, which is a function of voltage and/or current.

A method according to the embodiments of the invention may beimplemented on a computer as a computer implemented method. Executablecode for a method according to the invention may be stored on a computerprogram medium. Examples of computer program media include memorydevices, optical storage devices, integrated circuits, servers, onlinesoftware, etc. Accordingly, a white-box system may include a computerimplementing a white-box computer program. Such system, may also includeother hardware elements including storage, network interface fortransmission of data with external systems as well as among elements ofthe white-box system.

In an embodiment of the invention, the computer program may includecomputer program code adapted to perform all the steps of a methodaccording to the invention when the computer program is run on acomputer. Preferably, the computer program is embodied on anon-transitory computer readable medium.

Any combination of specific software running on a processor to implementthe embodiments of the invention, constitute a specific dedicatedmachine.

As used herein, the term “non-transitory machine-readable storagemedium” will be understood to exclude a transitory propagation signalbut to include all forms of volatile and non-volatile memory. Further,as used herein, the term “processor” will be understood to encompass avariety of devices such as microprocessors, field-programmable gatearrays (FPGAs), application-specific integrated circuits (ASICs), andother similar processing devices. When software is implemented on theprocessor, the combination becomes a single specific machine.

It should be appreciated by those skilled in the art that any blockdiagrams herein represent conceptual views of illustrative circuitryembodying the principles of the invention.

Although the various exemplary embodiments have been described in detailwith particular reference to certain exemplary aspects thereof, itshould be understood that the invention is capable of other embodimentsand its details are capable of modifications in various obviousrespects. As is readily apparent to those skilled in the art, variationsand modifications can be effected while remaining within the spirit andscope of the invention. Accordingly, the foregoing disclosure,description, and figures are for illustrative purposes only and do notin any way limit the invention, which is defined only by the claims.

What is claimed is:
 1. A power converter comprising: a dual outputresonant converter including a first output, a second output, a commonmode control input, and a differential mode control input, wherein avoltage/current at the first output and a voltage/current at the secondoutput are controlled in response to a common mode control signalreceived at the common mode control input and a differential modecontrol signal received at the differential mode control input; and adual output controller including a first error signal input, a seconderror signal input, a common mode control output, and a differentialmode control output, wherein the dual output controller is configured togenerate the common mode control signal and the differential modecontrol signal in response to a first error signal received at the firsterror signal input and a second error signal received at the seconderror signal input, wherein the first error signal is a function of thevoltage/current at the first output and the second error signal is afunction of the voltage/current at the second output, and wherein thecommon mode control signal is output from the common mode control outputand the differential mode control signal is output from the differentialmode control output, wherein the differential mode control signal islimited by a differential mode signal limiting circuit; furthercomprising a clamping circuit configured to clamp the common modecontrol signal to a range of values; wherein the clamping circuitfurther comprises, a power detector configured to produce an indicationsignal when an output power of the first output approaches zero; and alimit detector configured to receive the common mode control signal andthe indication signal to produce a limited common mode signal based uponthe range of values.
 2. The power converter of claim 1, wherein thedifferential mode signal limiting circuit includes: a first maximumdetector receiving a scaled first error signal at a first input and afirst differential mode signal limit at a second input, wherein thefirst maximum detector is configured to produce an output that is amaximum value received at the first input and the second input; a secondmaximum detector receiving a scaled second error signal at a third inputand a second differential mode signal limit at a fourth input, whereinthe second maximum detector is configured to produce an output that is amaximum value received at the third input and the fourth input; a firstadder configured to produce an output that is an addition of the firstmaximum detector output and the second maximum detector output; and asecond adder configured to produce the differential mode control signalby subtracting the output of the first adder from a differential modesignal maximum value Vdm_max.
 3. The power converter of claim 2, whereinthe differential mode signal maximum value is scaled by a value kwherein 1<k<1.1.
 4. The power converter of claim 2, wherein the firstdifferential mode signal limit is calculated as Vdm_max*alpha/m, whereinalpha and m are scale factors such that 0<alpha<1 and 1<m<1.1.
 5. Thepower converter of claim 1, wherein the common mode control signal isgenerated using a feedback loop that uses a desired delta power signalbased upon the first error signal and the second error signal and adelta power signal that is a function of the difference in output powerat the first output and the second output.
 6. The power converter ofclaim 5, wherein the dual output controller is configured to generatethe common mode control signal and the differential mode control signalin response to the first error signal and the second error signal byprecalculating a control variable matrix and generating the common modecontrol signal and the differential mode control signal as a function ofthe first and second error signals and the control variable matrix. 7.The power converter of claim 6, wherein the control variable matrixincludes coefficients k11, k12, k21, and k22, wherein the desired deltapower signal and the differential mode control signal are generated as:desired_delta_power=first error signal·k21+second error signal·k22; andVdm(differential mode control signal)=first error signal·k11+seconderror signal·k12.
 8. The power converter of claim 1, wherein the powerdetector includes a reference input configured to receive a referencesignal and a voltage/current input that receives a voltage/currentsignal from the first output, wherein the indication signal producedbased upon the reference signal and the voltage/current signal from thefirst output.
 9. The power converter of claim 1, wherein the powerdetector includes an auxiliary voltage input configured to receive anauxiliary voltage signal from an auxiliary coil coupled to a secondaryof the dual output resonant converter, wherein the indication signal isproduced based upon the auxiliary voltage signal.
 10. The powerconverter of claim 1, wherein the power detector includes an auxiliaryvoltage input configured to receive an auxiliary voltage signal from anauxiliary coil coupled to a secondary of the dual output resonantconverter and a switching node voltage input that receives a switchingnode voltage signal from a switching node in the dual output resonantconverter, wherein the indication signal is produced based upon theauxiliary voltage signal and the switching node voltage signalindicating a beginning of a continuous conduction mode (CCM) ofoperation of the dual output resonant converted.
 11. A power convertercomprising: a dual output resonant converter including a first output, asecond output, a common mode control input, and a differential modecontrol input, wherein a voltage/current at the first output and avoltage/current at the second output are controlled in response to acommon mode control signal received at the common mode control input anda differential mode control signal received at the differential modecontrol input; and a dual output controller including a first errorsignal input, a second error signal input, a common mode control output,and a differential mode control output, wherein the dual outputcontroller is configured to generate the common mode control signal andthe differential mode control signal in response to a first error signalreceived at the first error signal input and a second error signalreceived at the second error signal input, wherein the first errorsignal is a function of the voltage/current at the first output and thesecond error signal is a function of the voltage/current at the secondoutput, and wherein the common mode control signal is output from thecommon mode control output and the differential mode control signal isoutput from the differential mode control output, wherein thedifferential mode control signal is limited by a differential modesignal limiting circuit; wherein the differential mode signal limitingcircuit includes: a first maximum detector receiving a scaled firsterror signal at a first input and a first differential mode signal limitat a second input, wherein the first maximum detector is configured toproduce an output that is a maximum value received at the first inputand the second input; a second maximum detector receiving a scaledsecond error signal at a third input and a second differential modesignal limit at a fourth input, wherein the second maximum detector isconfigured to produce an output that is a maximum value received at thethird input and the fourth input; a first adder configured to produce anoutput that is an addition of the first maximum detector output and thesecond maximum detector output; and a second adder configured to producethe differential mode control signal by subtracting the output of thefirst adder from a differential mode signal maximum value Vdm_max. 12.The power converter of claim 11, wherein the differential mode signalmaximum value is scaled by a value k wherein 1<k<1.1.
 13. The powerconverter of claim 11, wherein the first differential mode signal limitis calculated as Vdm_max*alpha/m, wherein alpha and m are scale factorssuch that 0<alpha<1 and 1<m<1.1.
 14. The power converter of claim 11,wherein the common mode control signal s generated using a feedback loopthat uses a desired delta power signal based upon the first error signaland the second error signal and a delta power signal that is a functionof the difference in output power at the first output and the secondoutput.
 15. The power converter of claim 14, wherein the dual outputcontroller is configured to generate the common mode control signal andthe differential mode control signal in response to the first errorsignal and the second error signal by precalculating a control variablematrix and generating the common mode control signal and thedifferential mode control signal as a function of the first and seconderror signals and the control variable matrix.
 16. The power converterof claim 15, wherein the control variable matrix includes coefficientsk11, k12, k21, and k22, wherein the desired delta power signal and thedifferential mode control signal are generated as:desired_delta_power=first error signal·k21+second error signal·k22; andVdm(differential mode control signal)=first error signal·k11+seconderror signal·k12.